PROCEEDINGS OF THE 9TH CANADIAN WORKSHOP ON INFORMATION THEORY, MONTREAL, QUEBEC, JUNE 5 - 8, 2005

A HARDWARE METHOD FOR REAL-TIME SNR ESTIMATION FOR M-PSK USING A SYMBOL SYNCHRONIZATION LOCK METRIC Yair Linn University of British Columbia 2111 Lower Mall, Room 1109, Vancouver, BC, Canada V6T-1Z4 email: [email protected] ABSTRACT

2

1

SIGNAL AND RECEIVER DEFINITIONS

The baseband M-PSK signal before modulation is defined as

In this paper we present and analyze a method for channel SNR (Signal to Noise Ratio) estimation for M-PSK, which is based upon the exploitation of the statistics of a symbol synchronization lock metric. The analysis pertains to AWGN (Additive White Gaussian Noise) channels in systems whose baseband pulse shape is Square-Root Raised Cosine (SRRC) or rectangular. It is shown that, in terms of latency, the proposed SNR estimation method often offers an improvement of several orders of magnitude as compared to estimation via the SER (Symbol Error Rate) or the BER (Bit Error Rate). The proposed method is also very simple to implement in hardware and is resistant to fading and to imperfections in the operations of the AGC (Automatic Gain Control) circuit.

m(t ) 



∑ exp ( jφ ) p(t − rT )

where

r

1/ T

is the

r =−∞

symbol

φr  2π ⋅ mr M + χ M ⋅ π M

rate,

mr ∈ {0,1,..., M − 1} ,

,

χ M  {1 if M ≠ 2,

and

0 if M = 2} . We shall first assume that the baseband pulse shape p (t ) is Square-Root Raised-Cosine (SRRC), that is ([7] eq. 68.15):

p ( t ) = 4α

INTRODUCTION

In a growing number of contemporary wireless communications systems, estimation of the channel SNR has become one of the most important tasks that the receiver must handle. In addition to the obvious utility of SNR estimates in so far as monitoring the receiver’s status, in many systems SNR estimates play a much more important role, namely - they play a part in the actual demodulation process. For example, diversity reception systems use SNR estimates in order to assign relative weights to the data obtained from the various antennas and/or receivers that make up the diversity scheme[4]. In various adaptive schemes, the data and/or coding rates are altered according to the SNR. Finally, receivers which use turbo codes (or other SNR-optimizable codes) can make use of an SNR estimate to increase their coding gain[6]. An immediate conclusion that can be drawn from the use of SNR estimates in the reception process is that those estimates must be generated in real-time; using an outdated SNR estimate can do much more harm than good. In this paper, we present an SNR estimation method that can provide accurate real-time SNR estimates using few hardware resources. The method is based on the exploitation of the statistics of a symbol synchronization lock detector which has been described for BPSK and QPSK in [1] and [2]. Other advantages of the proposed method include the fact that it is NDA (Non Data Aided) and that it requires only two samples per symbol which correspond to those needed for the Gardner Timing Error Detector (TED) [5] and related detectors ([10], [11]).

cos ((1 + α )π t T ) +

sin ((1 − α )π t T )

4α t T π T (1 − 16α 2 t 2 T 2 )

(1)

yielding the raised-cosine channel response of ([8] eq. 2.2.10):

g (t )  p ( t ) ⊗ x ( t ) =

sin (π t / T ) cos (πα t T ) ⋅ 2 πt /T 1 − 4α 2 t 2 T

(2)

where 0 ≤ α ≤ 1 is the “rolloff” factor and x (t ) is the matched filter (see Fig. 1). In Section 7 we shall also briefly discuss the case of rectangular baseband pulses. At the input of demodulator the IF signal is[4] the I -Q

sm (t ) = Re[ m(t ) exp( jω i t + jθ i )]

and that signal is

corrupted by AWGN. An M-PSK coherent receiver has the structure shown in Fig. 1, where: 1. 1/ TS 2.

= 2/T

is the sample rate.

n(t ) ~ N (0, N 0W )

bandpass IF filter before the 3.

KI

KQ

and

where

W

is the width of the

I - Q demodulator (not shown).

are the physical gains associated with the

circuit. These gains are a slow function of time controlled by the AGC circuit, which aims to ensure that the dynamic range of the samplers is utilized yet the samplers are not saturated. 4. We assume a locked carrier, i.e. ∆ω = 0 and

θ o ∈ {θ i + 2π k / M k = 0,1,..., M − 1} , where 2π k / M is

the inherent M-fold carrier synchronization ambiguity. 5. The matched filter x (t ) = p (−t ) is assumed ideal.

This work was supported by an NSERC (National Sciences and Engineering Research Council of Canada) Postgraduate Scholarship.

3 Denote

247

REVIEW OF LOCK DETECTOR the

even

samples

of

the

channels

as

PROCEEDINGS OF THE 9TH CANADIAN WORKSHOP ON INFORMATION THEORY, MONTREAL, QUEBEC, JUNE 5 - 8, 2005

Re m(t) ⋅ exp ( jω i ⋅ t + jθ i ) + n(t)

KI

x(t )

I(t)

2 cos(ω i ⋅t + ∆ ω ⋅t +θ o ) Local Carrier

IF Input

Carrier Sync./ Local Carrier Generation

−2sin(ω i ⋅t +∆ ω ⋅t +θ o )

90o

x(t ) KQ Sampling Clock 1/TS=2/T

Symbol Timing Error Detector (e.g. Gardner TED)

Symbol Sync./ Sampling Clock Generation

Q(t)

Symbol Sync. Lock Detector

Fig. 1. Simplified structure of an M-PSK receiver.

I e (n)  I (t ) t =2nT

and

S

odd

samples

Qe (n)  Q(t) t=2nT

S

, and the

Io (n)  I (t ) t =(2n+1)T

as

S

Qo (n)  Q(t) t=(2n+1)T

S

and

. We can then compute a symbol

synchronization lock detection metric ([1], [2]):

  I e 2 (n) − I o 2 (n)       2 2 1 N   I e ( n) + I o ( n)   sN  ∑   (3) 2 2 2 N n =− N +1   − Q ( n ) Q ( n ) e o  +χM    2 2  + Q ( n ) Q ( n ) o  e   (note: χ M was defined in Section 2). The lock detectors

Fig. 2.

described in [1] and [2] are suitable for use in conjunction with the Gardner TED[5] or related detectors ([10],[11]), since the latters require the same sample sets. It is further shown[2] that the

distribution

of

sN locked ~ N ( f M ,α ( ES /N 0 ) , σ 2 )

s N unlocked ~ N (0 , σ 2 ) , being the expected value of

sN

with

for a given

sN

For any

SNR ESTIMATION FROM SN

λ = f M−1,α (sN ) , namely the estimation of the ES / N 0 ratio from the value of sN . The ES / N 0 estimate is usually

f M ,α ( ES / N 0 )

desired in units of dB, as follows:

M and α and

λ d B = 1 0 ⋅ lo g 1 0

(f

−1 M ,α

(sN ))

(5)

Graphs of (5) are given in Fig. 2, which was obtained via stochastic simulations (i.e. equation (3) with simulated inputs).

M and any α the function f M ,α (•) is strictly

4.2 Hardware Implementation Fig. 3 shows an efficient hardware structure for (5). Since it is

1

monotonic , which means that an estimate of the channel −1

ES / N 0 can be obtained via f M ,α ( sN ) when the receiver is

seen

locked. The investigation of this estimation technique is the focus of this paper.

e

(I

from

2

(Q

e

1

4

sN . Top: M=2, 4. Bottom: M=8, 16.

We are interested in investigating the inverse relation

and

(4)

2 3(1− χ M ) N + N

vs.

4.1 Mathematical Formulation of Estimation Method

is

where ([1],[2]):

σ2 ≤

λdB

See [1],[2] for a demonstration of this for BPSK and QPSK.

inspection

(n) − I o (n) ) ( I e ( n) + I o ( n) ) ≤ 1 2

2

2

2

(n) − Qo 2 (n) ) (Qe 2 (n) + Qo 2 ( n) ) ≤ 1 ,

that and and

since a small dynamic range is needed for (5) (see Fig. 2), the

248

Qo (n)

Qe 2 ( n ) − Qo 2 ( n ) Qe 2 ( n ) + Qo 2 ( n )

Integrate and Dump Averager sum 2N samples and disregard lower log 2 (2N) bits

sN

Lookup Table:

sN → 10log10 ( f

−1 M ,α

O ut pu t

Lookup Table:

Output

I e 2 (n) − I o 2 (n) I e 2 ( n) + I o 2 ( n)

This branch for all M

A dd r es s

Qe (n)

Lookup Table:

Output

I o (n)

Address

I e (n)

Address

PROCEEDINGS OF THE 9TH CANADIAN WORKSHOP ON INFORMATION THEORY, MONTREAL, QUEBEC, JUNE 5 - 8, 2005

(sN ) )

λdB

This branch for M>2 only

Fig. 3. Efficient hardware implementation of the proposed SNR estimation method. lookup tables in Fig. 3 can be implemented as small fixed-point lookup tables in hardware. Hence, this ES / N 0 estimation

 E  E  f M ,α  (1+ r1 ) S  − f M ,α  S  N0    N0 

method is quite suitable for realization within an FPGA or ASIC. Observe how in Fig. 3 outright division by 2N is avoided, where N is assumed to be a power of 2. Moreover, if

sN

relation E [s N ] = f M ,α ( ES / N 0 ) :

   E  E P  f M ,α  (1 − r2 ) S  < s N < f M ,α  (1 + r1 ) S N0  N0   

is already generated for lock detection, then SNR

estimation necessitates only an additional small lookup table that computes λ dB , a trivial addendum. From (4) it is apparent that to achieve a more accurate value of

Since ([1],[2])

should be increased until

sN

is Gaussian, we conclude from (8) that in

 sN − E [sN ]  y <   Var ( s ) Var ( sN )  N 

P

Mathematically, the question is: What is the minimal value of N that is needed so that the following will hold?

  y = erf  >C  2 Var ( s )  N  

P f M−1,α ( s N ) − ( ES N 0 ) < tol > C

with erf ( x ) =

)

(6)

tol is the estimation tolerance and C is the confidence. Assume tol is in units of dB. We define the constants

2

π

x

∫e

−t

2

dt . Using (4) we solve the last

inequality to yield:

r1 = (10tol /10 − 1)

 erf −1 (C )  8 2N >   3(1 − χ M ) + 1  y 

f M ,α (•)

r2 = (1 − 10 − tol /10 ) .

Since

 >C 

(7)

   E  E  ⇔ P fM ,α  (1− r2 ) S  < sN < fM ,α  (1+ r1 ) S   > C N0  N0     

Define:

  E  E  y  m in  f M ,α  (1 − r2 ) S  − f M ,α  S  N0  N0 

2

(10)

To recap, we have just shown that choosing N which complies with (10) ensures the fulfillment of (6). Note that eq. (10) is expressed in terms of 2N (not simply of N ) since 2N is the number of symbol intervals used in computation of the lock metric (see (3)). Having now found a quantitative measure of the proposed SNR estimation method’s performance, let us now compare it to another popular estimation method - namely, SNR estimation via measurement of the Symbol Error Rate (SER).

is monotonically increasing we can rewrite (6) as:

 E E E P  − r2 ⋅ S < f M−1,α ( sN ) − S < r1 ⋅ S N0 N0 N0 

(9)

0

where

and

(8)

order for (6) to be true it is sufficient to require that:

the lock detector’s variance drops to an acceptable range. Indeed, as we shall see, a good yardstick by which we can measure the efficacy of the proposed SNR estimation method is by obtaining the answer to the following question: What is the minimal value of N needed to achieve a desired tolerance in the estimation of the ES / N 0 , with a desired confidence.

(

   

 s − E [s ]  y N  ≥ P N <  Var ( s N )  Var s ( ) N  

4.3 Quantitative Study of the Estimation Method

sN , and by extension of λ dB , N

  . Then using the 

5

,

SNR ESTIMATION VIA THE SER

One of the most prevalent methods of SNR estimation found in contemporary wireless receivers is estimation via the measurement of the pre-decoder or post-decoder error rate. In

249

PROCEEDINGS OF THE 9TH CANADIAN WORKSHOP ON INFORMATION THEORY, MONTREAL, QUEBEC, JUNE 5 - 8, 2005

order to assess the efficacy of the estimation method proposed in this paper, we shall now quantitatively compare the proposed method to SNR estimation through measurement of the pre-decoder SER. Why the pre-decoder SER and not the post-decoder error rate? Since for coded transmission the postdecoder error rate is smaller (often by orders of magnitude) as compared to the pre-decoder SER, it follows that the number of symbols required to obtain an SNR estimate from the predecoder SER is a lower bound on the number of symbols needed for SNR estimation using the post-decoder error rate. Moreover, results derived from post-decoder error rates would be inexorably linked to the particular coding scheme employed, hence limiting their usability as a benchmark for comparison. Define the SER of a received M-PSK signal at a given SNR as (see [3] eq. (16), [4] eq. 5.2-56)

Pe = 1 − ∫

π /M

−π / M

where

p(∆φ ) ⋅ d (∆φ )  g M ( ES N 0 )

p ( ∆φ ) is the Rician phase distribution given by[3]:

p ( ∆φ ) 

 −E  1 exp  S  2π  N0 

tol = 1 .5 dB with a confidence of C = 95% .

 E  2 ES cos ( ∆φ ) exp  S ⋅ cos 2 ( ∆φ )  × 1 + N0  N0   cos ( ∆φ )





2 ES N0

e− x

2

−∞

U i as U i = {1

Define the auxiliary variables

Fig. 4. Comparison of the required number of symbol intervals necessary to estimate the E S / N 0 to

  /2 dx    ⇔ an

required to produce the SNR estimate, as given in (10), to the number of symbol intervals which is needed in order to attain a likewise accurate estimate from the SER, as given in (13). This is shown in Fig. 4. Since the curves in Fig. 4 are similar to the graphs in Fig. 4 in [3], we can reach many of the same conclusions regarding the estimation method presented in this paper. Indeed, the discussion that now follows closely mirrors the one found in [3] Section V. When we look at Fig. 4 we can conclude by inspection that the proposed SNR estimation method is particularly advantageous for use at high-SNR. For low SNR, it might appear from Fig. 4 that estimation via the SER provides better performance; however, this is not necessarily the case. To see why, it should be noted that Fig. 4 assumes that the SER is measured perfectly. If the data being sent is a “training sequence” (i.e., a known sequence of symbols) then this would be a good assumption; however, sending training sequences does not convey information and thus represents a waste of the channel’s capacity. Conversely, if the data being sent is unknown (i.e., not a training sequence) then the assumption that the SER measurement is accurate becomes a progressively worse assumption as the SNR decreases. This is because the only way to achieve a direct SER measurement on unknown data would be to use an Error Correction Decoder (ECD) and find the errors by comparing the corrected data stream to the received symbols[9]. This assumes that the ECD’s output data stream is error free - a bad assumption at low SNR. Moreover, the ECD might not even be able to attain or maintain lock, hence rendering the SER measurement impossible in the first

(11)

error

was detected in symbol i, 0 otherwise}. We then have P (U i = 1) = Pe and P (U i = 0) = 1 − Pe , and therefore

E [U i ] = Pe

and

measured SER as estimate the

Var (U i ) = Pe (1 − Pe ) . Define the

S (L) =

1 L

L

∑U

i

, then (see [3]) we can

i =1

− ES / N 0 via η = g M ( S ( L) ) . So as to 1

arrive at a quantity to compare with (10), we are interested in finding a value of L that will ensure that:

 E E E  P  −r2 ⋅ S < η − S < r1 ⋅ S  > C N0 N0 N0  

(12)

In [3] it is shown that it is sufficient to ensure that:

L > Pe (1 − Pe where

(see

[3])

)

2

place. In contrast, estimation via

(13)

sN

does not depend on the

data and does not require an ECD. Indeed, as already

{ ) − P }.

z  min gM ((1 − r2 ) ⋅ ES N0 ) − Pe ,

g M ( (1 + r1 ) ⋅ ES N 0 6

 1  z  −1 2  e r f ( C ) 

mentioned, the SNR estimate produced via

sN

can be

provided to the ECD in order to improve the latter’s coding gain (e.g. turbo codes[6]). Finally, an additional advantage of

e

estimation via

DISPLAY AND ANALYSIS OF RESULTS

sN

is that the SNR estimate is very insensitive

to fading and AGC circuit imperfections. This is a direct result

An evaluation of the efficacy of estimating the SNR via (5) may be obtained by comparing the number of symbol intervals

of the fact that

250

sN

is extremely insensitive to such effects,

PROCEEDINGS OF THE 9TH CANADIAN WORKSHOP ON INFORMATION THEORY, MONTREAL, QUEBEC, JUNE 5 - 8, 2005

owing to its independence from

KI

and

KQ

(see [1], [2]).

In summary, we can say the following: (a) Despite what might be inferred from Fig. 4 estimation via

sN

has numerous

advantages at low SNRs as well; (b) As Fig. 4 shows, the biggest performance advantage is obtained when BPSK or QPSK is the modulation method. The required sampling rate of 2 samples/symbol is obviously a disadvantage as compared to estimation via the SER or via the method in [3], both of which required only one sample per symbol. Still, if the sampling rate is already twice the data rate (e.g., if the Gardner TED[5] or related detectors ([10],[11]) are employed) then no sampling rate increase is required in order to produce λ d B . The main advantages of the method discussed here w.r.t. the method in [3] is that (a) the method in [3] requires that

KI

and

KQ

in Fig. 1 be

identical, and (b) it can be shown that the estimate

λ dB

Fig. 5.

is

λdB

vs.

sN

for rectangular baseband pulses.

less sensitive to residual DC components that may present at the outputs of the samplers. Moreover, due to the negligible hardware resources needed to generate λ d B , the latter

well suited for providing real-time SNR estimates.

estimator may be used in conjunction with the method of [3] in order to derive a more accurate estimate and/or reduce the number of symbol intervals needed.

[1] Y. Linn, “A symbol synchronization lock detector and SNR estimator for QPSK, with application to BPSK,” in Proc. 3rd IASTED Intl. Conf. on Wireless and Optical Communications (WOC 2003), Jul. 14-16, 2003, Banff, AB, Canada, pp. 506-514. [2] Y. Linn, “A self-normalizing symbol synchronization lock detector for QPSK and BPSK,” IEEE Trans. on Wireless Commun., to be published. [3] Y. Linn, “Quantitative analysis of a new method for realtime generation of SNR estimates for digital phase modulation signals,” IEEE Trans. Wireless Commun., vol. 3, no. 6, pp. 1984-1988, Nov. 2004. [4] J. G. Proakis, Digital Communications, 4th ed., NY: McGraw-Hill, 2001. [5] F. M. Gardner, “A BPSK/QPSK timing error detector for sampled receivers,” IEEE Trans. Commun., vol. COM-34, no. 5, pp. 423-429, May 1986. [6] T. A. Summers and S. G. Wilson, “SNR mismatch and online estimation in turbo decoding,” IEEE Trans. Commun., vol. 46, no. 4, pp. 421-423, Apr. 1998. [7] J. D. Gibson (editor), The Communications Handbook, 2nd ed. Boca Raton, FL: CRC Press, 2002. [8] U. Mengali and A. N. D’Andrea, Synchronization Techniques for Digital Receivers. NY: Plenum Press, 1997. [9] N. Celandroni, E. Ferro, F. Potorti, “Quality Estimation of PSK Modulated Signals,” IEEE Commun. Magazine, p. 50-55, Jul. 1997. [10] Y. Linn, “A new NDA timing error detector for BPSK and QPSK with an efficient hardware implementation for ASIC-based and FPGA-based wireless receivers,” in Proc. IEEE Intl. Symp. on Circuits and Systems (ISCAS 2004), May 23-26, 2004, Vancouver, BC, Canada, vol. 4, pp. 465-468. [11] Y. Linn, “Two new decision directed M-PSK timing error detectors,” to be published in Proc. 18th Canadian Conference on Electrical and Computer Engineering (CCECE 2005), May 1-4, 2005, Saskatoon, SK, Canada.

7

REFERENCES

M-PSK WHERE p(t) IS RECTANGULAR

The preceding analysis can also be easily adapted to the case of M-PSK with rectangular baseband pulses, namely

p (t ) =

{ 1/ T

first graph

λdB

}

for - T / 2 ≤ t ≤ T / 2 . To do this, we vs.

sN

for this case, as shown in Fig. 5 (where

f M , R ec ( E S / N 0 )  E [ s N ] As seen in that figure,

λdB

for rectangular

p (t ) ).

for rectangular pulses behaves

exactly the same as for SRRC pulses with α = 1. A related phenomenon was also observed in [2] regarding the curve of

E[sN ]

vs. the SNR. Using the preceding observations, we

conclude that the curves in Fig. 4 for α = 1 are also applicable to the rectangular pulse case. Moreover, for the same reason, the equations in Section 4.3 with α = 1 also apply to the rectangular pulse case.

8

CONCLUSIONS

An SNR estimation method for M-PSK was analyzed. The proposed technique has numerous attractive features, which include NDA estimation, no dependence upon the coding scheme, resilience to fading and to AGC imperfections, and a compact fixed-point hardware implementation. A quantitative comparison was undertaken vs. estimation via the SER, whereupon it was found that the proposed method often necessitates orders-of-magnitude less symbol intervals, particularly for the cases of BPSK and QPSK. Moreover, the proposed estimator can be used in conjunction with the estimator in [3] in order to further reduce the estimation latency and/or increase the estimation accuracy. Due to the aforementioned reasons, the proposed estimation method is

251

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